Motor driving apparatus and method for control of motor revolution

ABSTRACT

This invention provides a motor driving apparatus that made it possible to reduce torque ripples including those attributed to load variation of the motor and an associated method for control of motor revolution. An output stage to a multiphase DC motor is comprised of power elements to supply output voltages to multiphase coils and a predriver to supply drive voltages to the power elements. A resistor means detects a current flowing through the power elements. A supply current detector detects a voltage signal produced across the resistor means as a supply current, using a high-speed ADC and a moving average filter. An output controller generates a PWM signal with a frequency lower than the frequency of the high-speed ADC so that the current detected by the supply current detector conforms to a current signal indicating a motor revolving speed and transfers the PWM signal to the output stage.

CROSS-REFERENCE TO RELATED APPLICATIONS

This is a continuation application of U.S. application Ser. No.12/138,103, filed Jun. 12, 2008, now allowed, the contents of which arehereby incorporated by reference into this application.

The disclosure of Japanese Patent Application No. 2007-216610 filed onAug. 23, 2007 including the specification, drawings and abstract isincorporated herein by reference in its entirety.

BACKGROUND OF THE INVENTION

The present invention relates to a motor driving apparatus and a methodfor control of motor revolution and relates to a technique that iseffectively used as a revolution control technique for a three-phasespindle motor of a hard disk drive (HDD), for example.

For driving the spindle motor of an HDD system, a soft switching methodis used in which PWM is performed for two phases duringcurrent-conducting phase switching to suppress a steep change incurrent. In order to rotate a motor, detecting the motor position isneeded. As a sensor-less method, it is known that the back electromotiveforce (BEMF) of the motor is detected during a non-conducting periodfollowing the current-conducting phase switching. Detecting zerocrossing of an inductive load current is described in JapaneseUnexamined Patent Publication No. Hei 10 (1998)-341588. A motor drivingdevice and an integrated circuit device for motor driving adapted toreduce a variation in motor running torque are described in JapaneseUnexamined Patent Publication No. 2005-102447.

The above Patent Document 2 was previously proposed by the presentinventors. To solve problems in the above Patent Document 1 and thelike, in Patent Document 2, output voltages are supplied to three-phasecoils by a three-phase DC motor output stage including a predriver forsupplying drive voltages to power MOSFETs. Current zero crossingdetection is performed by monitoring whether the gate-source voltage ofeach of the power MOSFETs is equal to or greater than a predeterminedvoltage. Output of the current zero crossing detection is used for PLLcontrol whereby the timing of current-conducting phase switching iscontrolled and a drive voltage with 180 degree (deg) conduction iscreated. The three-phase DC motor output stage including the predriverfor supplying drive voltages to the power MOSFETs supplies the outputvoltages to the three-phase coils. In particular, it generatessinusoidal drive voltages with a waveform pattern in which a down hookfor a lowest voltage phase which corresponds to GND and an up hook for ahighest voltage phase which corresponds to power supply alternate perelectric angle of 60 degree, the waveform being represented by acombination of linear functions. Thereby, sinusoidal currents flowthrough the three-phase coils.

[Patent Document 1]

-   Japanese Unexamined Patent Publication No. Hei 10 (1998)-341588

[Patent Document 2]

-   Japanese Unexamined Patent Publication No. 2005-102447

SUMMARY OF THE INVENTION

In the technique of the above Patent Document 2, as is depicted in FIG.12, in order to drive the three-phase DC motor with a proper torque, amotor drive current detected by a DC shunt resistor Rnf are used, PWMduty is controlled so that this current conforms to a current command,and PWMCLK is output. The motor drive current detected by the DC shuntresistor Rnf is amplified by a sense amplifier SA and converted into adigital value by an analog/digital converter ADC. In a current errordetector, a current error is obtained from a difference between thedetected motor drive current value and the current command (SPNCRNT).Then, a PWM duty (Duty) is determined by a compensator.

By an output controller that executes PWM modulation, each phase isdriven by a PWMCLK depending on the PWM duty. Supposing the case ofconstant torque driving, ripples will be observed in the currentdetected by the DC shunt resistor Rnf every electric angle of 60 deg,because a voltage pattern change occurs per electric angle of 60 deg forthe sinusoidal motor drive current. Consequently, if the current commandfor DC is referred to as is in the current error detector, currentdetection errors will occur periodically and ripples will be introducedin the PWM duty. Even if it is tried to generate sinusoidal drivevoltages for constant torque driving, exactly sinusoidal voltages cannotbe produced due to the influence of the ripples. Hence, when constanttorque driving is intended, anticipated ripples by the current detectionare added to the current command for DC so that current detection errorsdo not occur. Thereby, it becomes possible to perform current controlaccurately and produce exactly sinusoidal voltages even if there areripples in the detected current.

FIG. 13 shows waveforms observed during an operation of detecting aphase current, when a Nyquist ADC is used as the analog/digitalconverter ADC. By PWM modulating the sinusoidal voltages for threephases, the PWM duties of the waveforms of the U-phase, V-phase, andW-phase voltages change slightly over time. Due to this PWM operation,the voltage appearing across the DC shunt resistor Rnf changes dependingon the on/off states of the three phases. For example, if the U phase ison at low level and the V and W phases are on at high level, a currentflowing in the U phase appears in the voltage across the DC shuntresistor Rnf. Because the Nyquist ADC samples at the center of a PWMcycle, a current in one of the three phases is detected. This results inthat ripples are observed in the ADC output every electric angle of 60deg.

In Patent Document 2, however, a situation where the loads of the coilsof the motor are uneven is not assumed. Even if such assumption is made,it is impossible to add ripples to counteract varying individual coilloads of the motor and, therefore, the influence of the ripples cannotbe suppressed. Increased recording density of HDD units requires afurther reduction in vibration during motor revolution. It is,therefore, very important to enable driving the motor with reducedtorque ripples in spite of load variation of the spindle motor.

An object of the invention is to provide a motor driving apparatus thatis simple in structure and has high performance and high functionalityand an associated method for control of motor revolution. Another objectof the invention is to provide a motor driving apparatus that made itpossible to reduce torque ripples including those attributed to loadvariation of the motor and an associated method for control of motorrevolution. The above-noted objects and other objects and novel featuresof the invention will become apparent from the description of thepresent specification and the accompanying drawings.

One embodiment of the invention disclosed herein is briefly describedbelow. A motor driving apparatus includes the following means. An outputstage to a multiphase DC motor is comprised of power elements to supplyoutput voltages to multiphase coils and a predriver to supply drivevoltages to the power elements. A resistor means detects a currentflowing through the power elements. A supply current detector detects avoltage signal produced across the resistor means as a supply current,using a high-speed ADC and a moving average filter. An output controllergenerates a PWM signal with a frequency lower than the frequency of thehigh-speed ADC so that the current detected by the supply currentdetector conforms to a current signal indicating a motor revolving speedand transfers the PWM signal to the output stage.

Another embodiment of the invention disclosed herein is brieflydescribed below. In the output stage to the multiphase DC motor, thepredriver generates drive voltages and the power elements driven by thedrive voltages supply output voltages to the multiphase coils. Theresistor means detects a current flowing through the power elements. Themotor revolving speed is controlled by driving the motor as follows:detecting a supply current from a voltage signal produced across theresistor means by using a high-speed ADC and a moving average filter;generating a PWM signal with a frequency lower than the frequency of thehigh-speed ADC so that the thus detected supply current conforms to acurrent command for controlling the motor revolving speed; andtransferring the PWM signal to the output stage.

During a steady revolution state of a motor, there is almost one-to-onecorrespondence between supply current and motor toque. By detecting thissupply current and using it for control of motor revolution, it becomespossible to reduce torque ripples including those attributed to loadvariation among motor coils.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 shows a block diagram of a current control block that performsconstant torque driving by 180 deg conduction in a motor drivingapparatus according to the invention.

FIGS. 2A through 2D show waveforms of applied voltages allowingsinusoidal currents to flow in the motor in the present invention.

FIG. 3 is a block diagram of one embodiment of the Δ−Σ ADC and themoving average filter in FIG. 1.

FIG. 4 shows a frequency characteristic of the moving average filter inFIG. 3.

FIG. 5 shows waveforms observed in an operation of detecting a supplycurrent using the Δ−Σ ADC in FIG. 1.

FIG. 6 shows one set of operating waveforms to explain the inventiondisclosed herein.

FIG. 7 shows another set of operating waveforms to explain the inventiondisclosed herein.

FIG. 8 shows one set of operating waveforms to explain the supplycurrent control according to the invention disclosed herein.

FIG. 9 shows another set of operating waveforms to explain the supplycurrent control according to the invention disclosed herein.

FIG. 10 is an overall block diagram of one embodiment of a motor drivingapparatus according to the present invention.

FIG. 11 is a block diagram of an example of configuration of hard diskequipment as a whole using a motor driving control circuit to which thepresent invention is applied.

FIG. 12 is a block diagram of an example of an arrangement for motordriving of prior art.

FIG. 13 shows waveforms observed in an operation of detecting a phasecurrent in the arrangement of FIG. 12.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

FIG. 1 shows a block diagram of a current control block that performsconstant torque driving by 180 deg conduction in a motor drivingapparatus according to the invention. To drive a three-phase DC motorwith a proper torque, the current control block uses current valuesdetected by a DC shunt resistor Rnf, controls the PWM duty so that thecurrent value conforms to a current command, and outputs PWMCLK. Themotor drive current detected by the DC shunt resistor Rnf is amplifiedby a sense amplifier SA and converted into a digital value at high speedby a digital/analog converter (hereinafter referred to as Δ−Σ ADC). TheΔ−Σ ADC output passes through a moving average filter with atransmission zero set at a PWM frequency, whereby PWM frequencycomponents are removed, and this output Ips is subjected to currenterror detection.

The output Ips of moving average filter is an average of the currentflowing across the DC shunt resistor Rnf and, therefore, the detectedcurrent value corresponds to a supply current. That is, the above Δ−ΣADC and moving average filter are constituent elements of a supplycurrent detector. In a current error detector, a current error isobtained from a difference between the output Ips of the moving averagefilter and the current command SPNCRNT. Then, a PWM duty PWMD isdetermined by a compensator. An output controller that executes PWMmodulation drives each phase by a PWM signal which depends on the PWMduty PWMD and waveform profiles for 180 deg conduction.

As is the case for the foregoing Patent Document 2, waveform profilesare created by a 180 deg conduction waveform profile generator. To thisprofile generator, a phase setting parameter and a ramp settingparameter of a waveform profile, which have been set in a register, areinput through a serial port. Waveform profiles for three phases aregenerated; that is, a PWM pattern and SP1, SP2 patterns are generated.Through a current control loop as described above, the current controlblock operates to make the supply current Ips constant and drives themotor.

FIGS. 2A through 2D show waveforms of applied voltages allowingsinusoidal currents to flow in the motor. Given that a lowest voltagephase is set to “0” (grounded to GND) for three-phase drive voltagesshown in FIG. 2A, the three-phase drive voltages will be as shown inFIG. 2B. Given that a highest voltage phase is set to a “power supply”(grounded to power supply) for the three-phase drive voltages, thesevoltages will be as shown in FIG. 2C. Here, if drive voltages are thosein which a state grounded to GND and a state grounded to power supplyalternate every electric angle of 60 deg as shown in FIG. 2D, agrounding point is alternately switched between GND and power supplyevery electric angle of 60 deg, while the voltages have up and downwaveforms that alternate every electric angle of 60 deg. Accordingly, inthe case of FIG. 2D, if drive voltages for an electric angle of 60 degcan be generated, three-phase sinusoidal voltages can be realized. Thewaveform profile generator generates profiles corresponding to suchwaveforms.

Voltage waveform patterns in which up and down waveforms alternate,which are generated when driving the motor with a constant torque, aregenerated by a controller for constant torque driving as describedabove; however, this controller is omitted in FIG. 1. If a switchingdrive with a constant current is performed to boost the toque just afterstarting the motor (constant torque driving is not performed) in theabove-mentioned arrangement shown in FIG. 12, the switches may beswitched (to the 0 position) using a signal SINENA, the driving circuitmay operate with a PWM duty obtained by current control as is, and thecurrent error detection may be switched to DC current control.

Power Pw of the spindle motor is expressed, using Ips for supplycurrent, Vps for supply voltage, ω for angular frequency of motor, T fortorque, Rm for coil resistance, Ispn for motor drive current, Psw forswitching loss by PWM, as follows:

Pw=Vps×Ips=T×ω+Ispn ² ×Rm+Psw  (1)

For current control to make supply current Ips constant, the currentcontrol block operates to make the right-hand member of the aboveequation (1) constant. In a steady running state of the motor, the firstterm (T×ω) of the right-hand member of the above equation is dominating.Hence, by the current control, toque T can be made constant. Thereby,driving the motor with reduced torque ripples in spite of load variationof the motor can be accomplished.

FIG. 3 shows a block diagram of one embodiment of the Δ−Σ ADC and themoving average filter in FIG. 1. For the Δ−Σ ADC, a one-bit secondaryΔ−Σ ADC with improved bit precision by high-speed sampling at, forexample, about 20 MHz, is used. A suitable Δ−Σ ADC configuration can beselected according to required bit precision and over-sampling ratio toPWM cycles.

The ADC output is first input to a moving average filter 1 forapproximately 64-time averaging. This is inserted to prevent the bitprecision from deteriorating by a loopback noise and an average numberof times is selected according to the bit precision required for currentdetection. An output signal of the moving average filter 1 is input to amoving average filter 2 synchronized with PWM cycles. Since PWM cyclesare generated by a counter, by using a counter reset signal DIF toupdate the moving average filter 2, PWM frequency components can beremoved completely.

As for a delay in the whole moving average filter, because of a smallnumber of averaging times in the first stage, the supply current Ips canbe detected in about a half of a PWM cycle. Because the moving averagefilter exists in the current control loop, it has only a small delay.Thus, the bandwidth for current control can be enhanced and the effectof suppressing supply current ripples, namely, torque ripples can beimproved.

FIG. 4 shows a frequency characteristic of the moving average filter inFIG. 3. The frequency characteristic is shown, assuming that the PWMfrequency of the moving average filter in FIG. 3 is 40 KHz. Because ofthe transmission zero of the PWM frequency, PWM frequency components (40KHz) can be removed completely. In this way, an average of currentflowing across the DC shunt resistor Rnf is obtained and PWM componentscan be excluded from the detected current value Ips.

FIG. 5 shows waveforms observed in an operation of detecting a supplycurrent using the Δ−Σ ADC in FIG. 1. By PWM modulating the sinusoidalvoltages for three phases, the PWM duties of the waveforms of theU-phase, V-phase, and W-phase voltages change slightly over time. Due tothe PWM operation, the voltage appearing across the DC shunt resistorRnf changes depending on the on/off states of the three phases. Forexample, if the U phase is on at low level and the V and W phases are onat high level, a current flowing in the U phase appears in the voltageacross the DC shunt resistor Rnf.

Since the Δ−Σ ADC operates at 20 MHz, if the PWM frequency is 40 KHz,the ADC operation is performed 500 times during one PWM cycle to acquirean ADC oversampled signal. Because the PWM frequency components areremoved completely by the moving average filter, there is no nipple inthe output of the moving average filter and the supply current can bedetected accurately, whereas ripples would be observed every electricangle of 60 deg in the above-mentioned case where a phase current isdetected using the Nyquist ADC as shown in FIG. 13. In the event thatripples occur in the torque due to load variation of the motor, theripples appear in the supply current Ips. Thus, current control isperformed to make the current Ips constant and the current control blockoperates to reduce the torque ripples. That is, in the above equation(1), because the supply voltage Vps is constant and T×ω>Ispn²×Rm+Psw,there is one-to-one correspondence between supply current Ips and toqueT. In other words, electric energy (Vps×Ips) supplied from the powersupply is converted into rotational energy (T×ω).

FIG. 6 shows one set of operating waveforms to explain the inventiondisclosed herein. This figure shows the operating waveforms created bycomputer simulation of the phase current control illustrated in FIG. 12.These waveforms are presented for the purpose of comparison with thesupply current control according to the present invention. This figureexplains current control for an instance where there is no loadvariation among the motor coils, in particular, there is no variation inback EMF. In the figure, U-phase voltage, U-phase back EMF, U-phasecurrent, supply current, torque, ADC output, and current flowing acrossthe DC shunt resistor Rnf are shown from top to down.

Each phase operates with a signal obtained by PMW modulation of drivevoltages corresponding to the waveform profiles shown in FIG. 2D. Thus,the U-phase voltage has a PWM waveform shown in FIG. 2D. Since theU-phase drive voltage is sinusoidal as noted above and the back EMF issinusoidal, the U-phase current is also a sinusoidal current. In thecase where there is no variation in back EMF, the current controloperates to reduce ripples in both the supply current and the toque. Thecurrent flowing across the DC shunt resistor Rnf is determined by thePWM state in each phase. As already noted, there are ripples everyelectric angle of 60 deg in the ADC output. In the above-mentionedarrangement for phase current control illustrated in FIG. 12, theseripples are corrected by SPNCRNT (broken line approximation).

FIG. 7 shows another set of operating waveforms to explain the inventiondisclosed herein. This figure also shows the operating waveforms createdby computer simulation of phase current control illustrated in FIG. 12for an instance where there is load variation among the motor coils. Inparticular, in the case that back EMF is small only in the U phase, thetorque constant in this phase decreases. In consequence, even if thephase current is made constant, the supply current and the torque arenot made constant. Ripples in the supply current and the torque causedegradation more significantly than in the case that there is novariation.

FIG. 8 shows one set of operating waveforms to explain the supplycurrent control according to the invention disclosed herein. This figureshows the operating waveforms created by computer simulation of thephase current control illustrated in FIG. 1. This figure explainscurrent control for an instance where there is no load variation amongthe motor coils, in particular, there is no variation in back EMF,corresponding to FIG. 6. In the figure, U-phase voltage, U-phase backEMF, U-phase current, supply current, torque, ADC output, and currentflowing across the DC shunt resistor Rnf are shown from top to down. Byperforming current control to make the moving average filter outputwhich corresponds to the supply current constant, ripples in the supplycurrent and toque can be suppressed.

FIG. 9 shows another set of operating waveforms to explain the supplycurrent control according to the invention disclosed herein. This figurealso shows the operating waveforms created by computer simulation of thephase current control illustrated in FIG. 1. This figure explainscurrent control for an instance where there is load variation among themotor coils, in particular, back EMF is small only in the U phase. It isseen that the supply current control can suppress ripples in the supplycurrent and torque as compared with the case of FIG. 7 by performingcurrent control to make the moving average filter output whichcorresponds to the supply current constant in spite of load variation ofthe motor. Ripple components due to load variation of the motor occur asthose that are n times as large as the back EMF. Owing to a small delayin the supply current detection of the present invention, the bandwidthfor current control can be enhanced and a favorable effect of reducingtorque ripples can be obtained.

FIG. 10 shows an overall block diagram of one embodiment of a motordriving apparatus according to the present invention. This embodiment isintended for constant torque driving of a three-phase motor as amultiphase motor by 180 deg conduction. The coils of the three-phasemotor are driven by pulse width modulation (PWM) signals from the outputstage comprised of power elements such as power MOSFETs M1-M6 and anoutput predriver. The output predriver operates, taking inputs of PWMmodulated signals UP, UN, VP, VN, WP, WN. These signals are produced inthe output controller by PWM modulation of sinusoidal drive voltagesgenerated by a drive voltage profile generator.

Spindle output voltages U, V, W are input to a selector SEL1 thatselects which phase in which BEMF is detected. A differential voltagebetween the SEL1 output and a neutral point CT of the motor is generatedby a preamplifier SA1. The differential voltage generated by thepreamplifier SA1 is filtered by a prefilter PFL. Then, the differentialvoltage is compared to a reference voltage Vref1 in a voltage comparatorCP 1 and a zero crossing of BEMF is detected.

Gate voltages UL, VL, WL of lower power MOSFETs M2, M4, M6 are input toa selector SEL2 that selects which phase in which a current zerocrossing is detected. The output of the selector SEL1 is compared to areference voltage Vref3 in a voltage comparator CP3. The output of theselector SEL2 is sample-held by S/H during a crossing time, that is, thetransition period of an output voltage. A voltage comparator CP2compares the sample-held voltage and a reference voltage Vref2 and azero crossing of current is detected.

Of these zero crossing outputs, which is to be used is selected by aselector SEL3. The current zero crossing detected is used, if driving by180 deg conduction is performed; otherwise, the BEMF zero crossingdetected is used. This is intended to use a stable start mode by BEMFdetection when the motor runs at a low rate and its revolution isunstable.

The detected zero crossing which is output from the selector SEL3 isinput to the phase error detector PD. By a PLL control loop comprised ofa compensator PLL, a conduction timing counter CNT, the outputcontroller OUTC (PWM), the predriver stage PDC, the power MOSFETs M1-M6,and the motor, motor driving with a low torque jitter is accomplished.The PLL control loop has a loop gain adjusting function in the case ofusing the current zero crossing.

When the current zero crossing is used to detect a phase error, there isno need for a non-conducting period which is provided when BEMF isdetected and, thus, 180 deg conduction can be achieved. Motor drivingwith a virtually constant torque and with reduced torque ripples can beaccomplished with sinusoidal drive voltages. Meanwhile, currentdetection is performed using the DC shunt resistor Rnf. A voltagedetected by the DC shunt resistor Rnf is amplified by the senseamplifier SA1 and input to the Δ−Σ ADC (ADC2) that performs oversamplingat high speed. Output of the Δ−Σ ADC is input to the moving averagefilter FL with a transmission zero at the PWM frequency. A supplycurrent Ips from which PWM frequency components have been removedcompletely can be detected.

Output Ips of the moving average filter FL which corresponds to thesupply current is input to the current control error detector IED wherean error of the current as a difference from a current command SPNCRNTis calculated. By the current control loop comprised of the currentcontrol compensator IC, output controller OUTC (PWM), predriver stagePDV, power MOSFETs M1-M6, and the motor, control is performed to makethe supply current Ips constant. In the current control error detectorIED, an error as a difference from the current command SPNCRNT iscalculated. A detailed block diagram including the current controlcompensator IC and the output controller OUTC (PWM) is shown in FIG. 1.

For motor speed control, a PHASE signal that is a periodic signalproportional to the number of revolutions is output by the outputcontroller OUTC. This signal is compared to a target speed by amicrocomputer or microprocessor MPU. A current command SPNCRNT inaccordance with the speed is created and input through the serial portSPORT.

Interfacing with the microcomputer or microprocessor MPU is performed bythe serial port SPORT. Parameters PM1-PM5 are set by the register PREG.Parameters such as a current command (SPNCRNT), current control, PLL,drive voltage profiles for 180 deg conduction are set by the register.The apparatus further includes a COMSENS controller COMC which controlsstarting from the motor stop state as a spindle control system and asequencer LOG which controls internal operation. To the COMSENScontroller COMC, output of the prefilter PFL, after converted into adigital value by the ADC 1, is input. The PM1 is a parameter for COMSENScontrol. The PM2 is a parameter for loop gain adjustment. The PM3 is aparameter for current control. The PM 4 is a parameter for PLL control.

FIG. 11 shows a block diagram of one example of configuration of harddisk equipment as a whole, as an example of a magnetic disk systemincluding a spindle motor control unit using a motor driving controlcircuit to which the present invention is applied, a voice coil motorcontrol unit, and a magnetic head driving control unit. In FIG. 11,reference numeral 310 denotes a spindle motor that rotates a magneticdisk 300; reference numeral 320 denotes arms, each having a magnetichead HD at its leading end (a magnetic write head and a magnetic readhead are included); reference numeral 330 denotes a carriage on whichthe arms 320 are rotatably supported. The voice coil motor 340 moves thecarriage 330, thereby moving the magnetic heads, and the motor drivingcontrol circuit 410 performs servo control to align the center of amagnetic head to the center of a track.

The motor driving control circuit 410 a semiconductor integrated circuitin which a spindle motor driving control circuit having functions asdescribed in the foregoing embodiment and a voice coil motor drivingcontrol circuit that moves the magnetic heads in a radial direction ofthe disk are integrated. This circuit operates in accordance with acontrol signal supplied from a controller 420 and performs servo controlof the voice coil motor 340 and the spindle motor 310 to seek a desiredtrack and move the appropriate magnetic head to the desired track and tomake the relative speed of each magnetic head constant. In this case,the above-mentioned power MOSFETs M1-M6 may be external elements orbuilt in the semiconductor integrated circuit. Since elements that carrya large drive current are hard to incorporate into the semiconductorintegrated circuit, it is desirable to provide them as external elementsas above.

Reference numeral 430 denotes a read/write IC that amplifies a currentin accordance with a change in magnetism detected by the magnetic headHD and sends a read signal to a signal processing circuit (data channelprocessor) 440 as well as amplifies a write pulse signal from the signalprocessing circuit 440 and outputs a drive current for the magnetic headHD.

Reference numeral 450 denotes a hard disk controller that takes in readdata sent from the signal processing circuit 440 and performs errorcorrection as well as performs error correction and coding of write datafrom a host and outputs it to the signal processing circuit 440. Thesignal processing circuit 440 performs modulation/demodulationprocessing suitable for digital magnetic recording and signal processingsuch as waveform shaping taking magnetic recording characteristics intoaccount as well as reads positional information from a read signal bythe magnetic head HD.

Reference numeral 460 denotes an interface controller that transfersdata from the present system to an external device and vice versa andperforms transfer-related control. The hard disk controller 450 iscoupled via the interface controller to a host computer such as amicrocomputer within a personal computer main body. Reference numeral470 denotes a cache memory for a buffer that temporarily stores readdata which has been read at a high rate from a magnetic disk. A systemcontroller 420 comprising a microcomputer determines which operationmode based on a signal from the hard disk controller 450 and controlseach part of the system according to the operation mode as well ascalculates a sector position based on address information supplied fromthe hard disk controller 450.

In the foregoing embodiment, as described in the above-mentioned PatentDocument 2 also, a zero crossing of a current waveform is detected bydetermining whether a gate-source voltage Vgs is present in the outputpower MOSFETs during a transition period of an output voltage, duringwhich PWM operation is performed. By using the current zero crossing,180 deg conduction without an non-conducting period of the motor can beaccomplished. Also, phase error detection in PLL control utilizesdetection results depending on whether the gate-source voltage Vgs ispresent during an output transition period. As a phase error, by using aquantity proportional to a difference of the number of detected polaritychanges of a drive current during an error detection period, it ispossible to perform PLL control not affected by detection offsets anddrive current ripples which occur in actual operation. Further, supplycurrent control to counteract load variation of the motor isaccomplished and ripples in the motor driving torque can be reducedsignificantly.

By provision of the selector for selecting either BEMF zero crossing orcurrent zero crossing as motor position information, it is possible tostart the motor by driving mode by conventional BEMF detection duringunstable motor rotation at a low rate. By the gain adjustment functionincluded in the PLL control, it is possible to perform operation withthe same loop characteristics whether BEMF zero crossing or current zerocrossing is used and it is possible to suppress degradation by torquejitter or the like. Drive voltages have waveform patterns in which upand down waveforms that alternate every electric angle of 60 deg. Byrepresenting these patterns by linear approximation, highly accuratesinusoidal drive voltages can be obtained by a simple structure andmotor driving with a constant torque can be achieved.

In these patterns, as up and down waveforms alternate every electricangle of 60 deg, current waveforms can have a good up and downsymmetrical property. This can prevent second-order distortioncomponents of torque ripples. By provision of a phase setting registerand a ramp setting, drive voltage phases and drive voltage distortioncan be adjusted independently of the motor and it is possible to drivethe motor with an optimum torque. A method of adjusting drive voltagedistortion by changing the ramp of up and down waveforms that alternateevery electric angle of 60 deg makes it easy to insert a sixth-orderdistortion. Therefore, this method is effective in compensating theinfluence of distortion components (mainly, a fifth-order andseventh-order combination) that exist in BEMF. In drive current control,by superposing anticipated ripple components on waveforms by a currentcommand, current control can be performed without giving rise to acurrent detection error. Thereby, it is possible to attain sinusoidaldrive currents.

The motor driving apparatus to which the present invention is appliedaccomplishes 180 deg conduction without a non-conducting period. Bydetecting the supply current and controlling the motor revolution, it ispossible to reduce torque ripples against load variation of the motorand to also reduce noise and vibration of the motor. Owing to 180 degconduction, it is possible to reduce ripples in the supply current toHDD equipment and to couple more HDDs to the power supply. By the phaseerror detection method not affected by detection offsets and drivecurrent ripples, 180 deg conduction can be attained even for relativelysmall motors such as a motor for a smaller number of disks and a motorwith a low number of revolutions.

While the invention made by the present inventors has been describedspecifically based on its embodiments hereinbefore, it will beappreciated that the present invention is not limited to the describedembodiments and various modifications may be made without departing fromthe gist of the invention. For example, the high-speed ADC employed issolely required to perform sampling at a sufficiently high rate relativeto the PWM frequency and attain desired bit precision. Although aone-bit secondary Δ−Σ ADC is used in the described embodiment, thenumber of bits and order can be changed according to characteristics.The moving average filter employed is solely required to have thecapability of removing PWM frequency components completely and any othertype of filer may be used, provided it has a transmission zero at thePWM frequency.

To detect a current zero crossing for implementing 180 deg conduction,it is determined whether a gate-source voltage Vgs is present in thelower power MOSFETs M2, M4, and M6; alternately, it may be determinedwhether a gate-source voltage Vgs is present in the upper power MOSFETsM1, M3, and M5. In the latter case, it should be determined whether arespective voltage between terminals UU and U, terminals VU and V, andterminals WU and W of the output stage is larger than a thresholdvoltage Vth for the MOSFETs. The power MOSFETs may be built in theintegrated circuit for motor driving or provided as external terminals,as noted above.

The driving mode is not only 180 deg conduction without a non-conductingperiod; alternatively, a driving mode may include only a fractionalnon-conducting period (about 10 deg) for detecting a BEMF zero crossing.Even in the latter mode, the effect is obtained which reduces torqueripples against load variation of the motor by detecting the supplycurrent and controlling motor revolution. Further, motor positiondetection may be accomplished using a sensor such as a hall element, asan alternative to using a current zero crossing as described above. Thepower MOSFETs as the power elements may be replaced with other elementssuch as bipolar transistors.

The present invention can widely be used for motor driving apparatus andas the method for control of motor revolution.

1. A motor driving apparatus comprising: a multiphase DC motor includingmultiphase coils; an output stage including: power elements which iscoupled to the multiphase coils and which supplies output voltages tothe multiphase coils, and a predriver which is coupled to the powerelements and which supplies drive voltages to the power elements; aresistor which is coupled to the power elements and which detects acurrent flowing through the power elements; a supply current detectorwhich includes an ADC and a digital filter and which detects a voltagesignal produced across the resistor as a supply current, using the ADCand the digital filter; and an output controller which generates a PWMsignal with a frequency lower than a sampling frequency of the ADC sothat the current detected by the supply current detector conforms to acurrent command for controlling the motor revolving speed, and transfersthe PWM signal to the output stage.
 2. The motor driving apparatusaccording to claim 1, wherein the digital filter is a moving averagefilter with a transmission zero set at the frequency of the PWM signal.3. The motor driving apparatus according to claim 1, wherein the ADC isa D-S ADC that performs oversampling at a higher frequency than thefrequency of the PWM signal.
 4. The motor driving apparatus according toclaim 3, wherein the digital filter is a moving average filter with atransmission zero set at the frequency of the PWM signal.
 5. The motordriving apparatus according to claim 4, wherein the power elementsinclude: upper power elements coupled to a supply voltage and lowerpower elements coupled to a ground potential, and wherein the resistoris a DC shunt resistor disposed between a common connection node for thelower power elements and the ground potential.
 6. The motor drivingapparatus according to claim 4, further comprising: a current zerocrossing detector that monitors whether a respective voltage to driveeach of the power elements is equal to or greater than a predeterminedvoltage and detects a current zero crossing; and a profile generatorthat generates a drive voltage of 180 degree conduction, using theoutput of the current zero crossing detector for phase locked loopcontrol that controls timing of conduction phase switching, wherein theoutput controller receives the drive voltage generated by the profilegenerator and generates a PWM signal to be transferred to the outputstage.
 7. The motor driving apparatus according to claim 6, furthercomprising: a back EMF zero crossing detector that selects anon-conducting phase and detects a back EMF zero crossing; and aselector that selects the back EMF zero crossing detector during lowrevolution, when the motor revolution is unstable, and transfers theback EMF zero crossing to the phase locked loop control, or selects thecurrent zero crossing detector during high revolution at a higher numberof revolutions than the low revolution, when the motor revolution fallsin a stable region, and transfers the current zero crossing to the phaselocked loop control.
 8. The motor driving apparatus according to claim7, wherein the current zero crossing detector includes a comparator thatcompares a gate-source voltage in the lower power elements to areference voltage that is equal to or less than a threshold voltage ofthe lower power elements and greater than 0 V.
 9. The motor drivingapparatus according to claim 7, wherein phase error detection in thephase locked loop control uses as a phase error a quantity proportionalto a difference of the number of detected polarity changes of a drivecurrent during an error detection period.
 10. The motor drivingapparatus according to claim 9, wherein the output stage is the outputstage to a three-phase DC motor having three-phase coils, and whereinthe profile generator generates a sinusoidal drive voltage with awaveform pattern in which a down hook for a lowest voltage phase whichcorresponds to GND and an up hook for a highest voltage phase whichcorresponds to power supply alternate every electric angle of 60 degree,and the waveform is represented by linear approximation.
 11. A methodfor control of motor revolution in an arrangement comprising: an outputstage to a multiphase DC motor including power elements to supply outputvoltages to multiphase coils and a predriver to supply drive voltages tothe power elements; and a resistor for detecting a current flowingthrough the power elements; a supply current detector which includes anADC and a filter; the method comprising: detecting a voltage signalproduced across the resistor as a supply current by using the ADC andthe filter in the supply current detector; generating a PWM signal witha frequency lower than a sampling frequency of the ADC so that the thusdetected supply current conforms to a current command for controllingthe motor revolving speed; and transferring the PWM signal to the outputstage for control of the motor revolving speed, wherein the filter iswith a transmission zero set at the frequency of the PWM signal.
 12. Amotor driving apparatus comprising: an output stage to a multiphase DCmotor including power elements to supply output voltages to multiphasecoils and a predriver to supply drive voltages to the power elements; aresistor for detecting a current flowing through said power elements; asupply current detector that detects a voltage signal produced acrosssaid resistor means as a supply current, using an analog to digitalconverter performing oversampling operation and a digital filter; and anoutput controller that generates a PWM signal with a frequency lowerthan the sampling frequency of the analog to digital converter so thatthe current detected by the supply current detector conforms to acurrent command for controlling the motor revolving speed, and transfersthe PWM signal to the output stage, wherein the digital filter is with atransmission zero set at the frequency of the PWM signal.
 13. The motordriving apparatus according to claim 12, wherein the analog to digitalconverter is a D-S ADC that performs oversampling at a higher frequencythan the frequency of the PWM signal.
 14. The motor driving apparatusaccording to claim 13, wherein the digital filter is a moving averagefilter with a transmission zero set at the frequency of the PWM signal.